Switching voltage regulator operating without a discontinuous mode

ABSTRACT

A switching circuit operates with a first operating state and a second operating state. During the first operating state, the switching circuit allows a switching current to linearly increase. During the second operating state, the switching circuit allows the switching current to linearly decrease. A control circuit is coupled to the switching circuit for controlling the switching circuit to operate with the first operating state or the second operating state. A setting circuit generates a threshold signal for the control circuit to ensure that during the first operating state the switching current linearly increases to become higher than or equal to a current value set by the threshold signal. Thereby, the switching current is prevented from linearly decreasing to reverse polarity during the second operating state.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a switching voltage regulator and, more particularly, to a switching voltage regulator operating without a discontinuous mode for enhancing power delivery efficiency.

2. Description of the Related Art

Switching voltage regulators supply a required output current at a regulated output voltage to a load. Through controlling duty ratios of power transistors, the switching voltage regulator converts an unregulated input voltage source to a stable, desired output voltage. FIG. 1 is a circuit block diagram showing a conventional synchronous switching buck regulator with a current feedback control. As shown in FIG. 1, a high-side switch HS and a low-side switch LS are series-connected between an input voltage source V_(in) and a ground potential. An inductor L has one end connected to a common node CN between the high-side switch HS and the low-side switch LS, and the other end serving as an output terminal for supplying an output voltage V_(out) to a load R_(L). The output terminal may be additionally provided with an output capacitor C_(o) for filtering the output voltage V_(out). The high-side switch HS and the low-side switch LS are controlled by a high-side drive signal HD and a low-side drive signal LD generated from the switching logic circuit 10, respectively. With respect to the synchronous switching voltage regulator, the high-side switch HS and the low-side switch LS are operated out of phase. An oscillating circuit 11 outputs a pulse signal PU having a fixed frequency to the switching logic circuit 10. In the beginning of each switch cycle, the switching logic circuit 10 turns on the high-side switch HS and turns off the low-side switch LS in response to the pulse signal PU. As a result, the input voltage source V_(in) supplies energy to the inductor L for linearly increasing the current I_(L). Once the inductor current I_(L) reaches a upper limit set by a slope-compensated error signal V_(err2) between a voltage feedback signal V_(vfb) and a reference voltage signal V_(ref), a comparator 12 is triggered to output HIGH instead of LOW. In response to the trigger event of the comparator 12, the switching logic circuit 10 turns off the high-side switch HS and turns on the low-side switch LS. As a result, the inductor current I_(L) linearly decreases since the energy stored in the inductor L delivers to the load R_(L).

FIG. 2 is a timing chart showing waveforms of the inductor current I_(L) of the synchronous switching buck regulator of FIG. 1. As shown in FIG. 2, a curve 21 indicates a waveform of the inductor current I_(L) with respect to a continuous-mode operation of the switching voltage regulator. Within each switch period T_(S), the inductor current I_(L) changes like a triangular wave having a linearly increasing portion corresponding to an operating state that the high-side switch HS is turned on for allowing the input voltage source V_(in) to supply energy to the inductor L and a linearly decreasing portion corresponding to another operating state that the high-side switch HS is turned off for allowing the energy stored in the inductor L to deliver to the load R_(L).

A curve 22 indicates a waveform of the inductor current I_(L) with respect to a discontinuous-mode operation of the switching voltage regulator. At a time t1, a switch cycle begins and therefore the high-side switch HS is turned on for linearly increasing the inductor current I_(L). At a time t2, the inductor current I_(L) reaches a upper limit I_(peak2) set by the slope-compensated error signal V_(err2), causing the high-side switch HS to be turned off and the low-side switch LS to be turned on. Consequently, the inductor current I_(L) starts to linearly decrease. At a time t3, the inductor current I_(L) has already decreased to zero although the next switch cycle will not start until a time t4. In this case, the inductor current I_(L) is subjected to polarity reversal between times t3 and t4, i.e. the flowing direction of the inductor current I_(L) makes a change of 180 degrees. For this reason, the conventional synchronous switching buck regulator must be additionally provided with a current reversal detecting circuit 17, as shown in FIG. 1, for triggering the switching logic circuit 10 to turn off the low-side switch LS immediately after the inductor current I_(L) decreases to zero. Thereby, the inductor current I_(L) is prevented from reversing polarity to reduce the power delivery efficiency.

Even if the current reversal detecting circuit 17 is provided or the circuit topology is replaced with a non-synchronous switching type which uses a combination of the power transistor and a flywheel diode as the switching circuit, the inductor current I_(L) is effectively prevented from reversing polarity. However, as shown in FIG. 2, the inductor current I_(L) remains zero between the times t3 and t4 with respect to the discontinuous-mode operation. In this case, the output voltage V_(out) inevitably rings or fluctuates up and down to cause unfavorable high-frequency noise.

SUMMARY OF THE INVENTION

In view of the above-mentioned problems, an object of the present invention is to provide a switching voltage regulator capable of being adapted to operate with the continuous mode, thereby avoiding the drawbacks caused by the discontinuous mode.

It is observed that if half the peak current flowing through the inductor is just equal to the average current flowing through the inductor, the switching voltage regulator operates at a threshold operating state between the continuous mode and the discontinuous mode. At such a threshold operating state, the peak current flowing through the inductor may be referred to as a threshold peak current of the continuous mode. Therefore, the switching voltage regulator according to the present invention is provided with a threshold peak current setting circuit for generating a threshold signal representative of the threshold peak current of the continuous mode. When the peak current flowing through the inductor is higher than the threshold peak current, it is unnecessary to adjust the inductor current since the continuous mode has been established. However, when the peak current flowing through the inductor is detected to be lower than the threshold peak current, the inductor current needs to be raised such that the peak current is substantially equal to the threshold peak current, thereby ensuring that the switching voltage regulator operates with the continuous mode.

According to one aspect of the present invention, a switching voltage regulator is provided to include a switching circuit, a control circuit, and a setting circuit. The switching circuit operates with a first operating state and a second operating state. During the first operating state, the switching circuit allows a switching current to linearly increase. During the second operating state, the switching circuit allows the switching current to linearly decrease. The control circuit is coupled to the switching circuit for controlling the switching circuit to operate with the first operating state or the second operating state. The setting circuit generates a threshold signal such that in response to the threshold signal the control circuit ensures that the switching current linearly increases to become higher than or equal to a current value set by the threshold signal during the first operating state. Thereby, the switching current is prevented from linearly decreasing to reverse polarity during the second operating state.

According to another aspect of the present invention, a method of controlling a switching voltage regulator is provided to include the following steps. A switching circuit is controlled to operate with a first operating state for allowing a switching current to linearly increase. The switching circuit is controlled to operate with a second operating state for allowing the switching current to linearly decrease. In the step of controlling the switching circuit to operate with the first operating state, the switching current is ensured to become higher than or equal to a threshold current. Thereby, the switching current is prevented from linearly decreasing to reverse polarity in the step of controlling the switching circuit to operate with the second operating state.

According to still another aspect of the present invention, a threshold current setting circuit is provided to include a first circuit, a second circuit, and a third circuit. The first circuit generates a first current signal in response to a voltage signal. The first current signal is proportional to the voltage signal. The second circuit generates a second current signal in response to the first current signal and a periodic signal. The second current signal is representative of a periodically-varying current. The third circuit generates a third current signal representative of a predetermined current value. A combination of the first to the third current signals is used to approximately simulate a threshold current signal.

BRIEF DESCRIPTION OF THE DRAWINGS

The above-mentioned and other objects, features, and advantages of the present invention will become apparent with reference to the following descriptions and accompanying drawings, wherein:

FIG. 1 is a circuit block diagram showing a conventional synchronous switching buck regulator with a current feedback control;

FIG. 2 is a timing chart showing waveforms of the inductor current with respect to continuous and discontinuous modes;

FIG. 3 is a timing chart showing a waveform of the inductor current with respect to a threshold operating state;

FIG. 4 is a circuit block diagram showing a synchronous switching buck regulator according to the present invention;

FIG. 5 is a detailed circuit diagram showing a first example of a threshold peak current setting circuit according to the present invention;

FIG. 6 is a detailed circuit diagram showing a second example of a threshold peak current setting circuit according to the present invention; and

FIG. 7 is a circuit block diagram showing a synchronous switching boost regulator according to the present invention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

The preferred embodiments according to the present invention will be described in detail with reference to the drawings.

It is observed from FIG. 2 that the relationship between the peak current and the average current flowing through the inductor presents an essential difference between the continuous and discontinuous modes. More specifically, in the continuous mode indicated by the curve 21, the average current I_(ave1) flowing through the inductor L is higher than half the peak current I_(peak1) flowing through the inductor L. However, in the discontinuous mode indicated by the curve 22, the average current I_(ave2) flowing through the inductor L is lower than half the peak current I_(peak2) flowing through the inductor L because during one part of each switch period T_(S) the inductor current I_(L) is substantially equal to zero given that a current reversal preventing device is provided. Therefore, in order to avoid the discontinuous mode, half the peak current flowing through the inductor must be made lower than or equal to the average current flowing through the inductor. As shown in FIG. 3, a curve 30 indicates a waveform of the inductor current I_(L) that reduces to zero immediately before the end of each switch cycle with respect to the threshold operating state. Through analytical calculation, the threshold peak current I_(peak) _(—) _(TH) is found to be determined cooperatively by the duty ratio D, the switch period T_(S), the input voltage V_(in), and the inductance L, as expressed in the following equation (1): $\begin{matrix} {I_{peak\_ TH} = {\left( \frac{T_{s}}{L} \right) \cdot D \cdot \left( {1 - D} \right) \cdot V_{i\quad n}}} & (1) \end{matrix}$

Since the threshold peak current I_(peak) _(—) _(TH) indicates the minimum allowable peak current of the continuous mode, the voltage regulator is effectively prevented from operating with the discontinuous mode if the peak current flowing through the inductor is made higher than or equal to the threshold peak current I_(peak) _(—) _(TH).

FIG. 4 is a circuit block diagram showing a synchronous switching buck regulator according to the present invention. As shown in FIG. 4, a high-side switch HS and a low-side switch LS are series-connected between an input voltage source V_(in) and a ground potential. An inductor L has one end connected to the common node CN between the high-side switch HS and the low-side switch LS, and the other end serving as an output terminal for supplying an output voltage V_(out) to a load R_(L). The output terminal may be additionally provided with an output capacitor C_(o) for filtering the output voltage V_(out). The high-side switch HS and the low-side switch LS are controlled by a high-side drive signal HD and a low-side drive signal LD generated from a switching logic circuit 40, respectively. The switching logic circuit 40 has an SR latch 41 for supplying the high-side drive signal HD and the low-side drive signal LD from the inverting output terminal {overscore (Q)}. In the embodiment shown in FIG. 4, the high-side switch HS and the low-side switch LS are implemented by signals with the same phase because the high-side switch HS is implemented by a PMOS transistor and the low-side switch LS is implemented by an NMOS transistor.

The SR latch 41 has a set terminal S connected to an output terminal of an inverter 42. The inverter 42 has an input terminal connected to an oscillating circuit 11 for receiving a pulse signal PU. The SR latch 41 has a reset terminal R connected to an output terminal of an NAND gate 43. The NAND gate 43 has a first input terminal connected to an output terminal of a comparator 12 for receiving a first comparison result signal CR1. The comparator 12 has an inverting input terminal connected to an output terminal of a slope compensation circuit 13. The slope compensation circuit 13 has two input terminals connected to the oscillating circuit 11 and an output terminal of an error amplifier 14, respectively, for generating a slope-compensated error signal V_(err2) in accordance with a ramp signal RA output from the oscillating circuit 111 and the error signal V_(err1), between a voltage feedback signal V_(vfb) and a reference voltage signal V_(ref), output from the error amplifier 14. The error amplifier 14 has an inverting input terminal connected to an output terminal of a voltage feedback circuit 15 for receiving the voltage feedback signal V_(vfb) representative of the output voltage V_(out) of the switching voltage regulator. The error amplifier 14 has a non-inverting input terminal for receiving the reference voltage signal V_(ref). The comparator 12 has a non-inverting input terminal connected to a current feedback circuit 16 for receiving a current feedback signal V_(ifb) representative of the inductor current I_(L).

The NAND gate 43 has a second input terminal connected to an output terminal of a comparator 44 for receiving a second comparison result signal CR2. The comparator 44 has an inverting input terminal connected to a threshold peak current setting circuit 45 for receiving a threshold signal V_(peak) _(—) _(TH). The comparator 44 has a non-inverting input terminal connected to the current feedback circuit 16 for receiving the current feedback signal V_(ifb). The threshold signal V_(peak) _(—) _(TH) generated by the threshold peak current setting circuit 45 is representative of the threshold peak current I_(peak) _(—TH) calculated in accordance with the equation (1) described above.

Hereinafter is described in detail the operation of the synchronous switching buck regulator according to the present invention with reference to FIG. 4. The pulse signal PU having a period T_(S) generated by the oscillating circuit 11 is supplied through the inverter 42 to the set terminal S of the SR latch 41. Since the SR latch 41 is a negative-going trigger circuit, the rising edge of the pulse signal PU after inverted triggers the SR latch 41 such that the inverting output terminal {overscore (Q)} supplies the high-side drive signal HD and the low-side drive signal LD with the LOW level for staring a switch cycle. The high-side drive signal HD with the LOW level turns on the high-side switch HS while the low-side drive signal LD with the LOW level turns off the low-side switch LS. As a result the input voltage source V_(in) supplies energy to the inductor L for linearly increasing the inductor current I_(L). The current feedback circuit 16 detects the inductor current I_(L) and then generates the current feedback signal V_(ifb) representative of the inductor current I_(L). The current feedback signal V_(ifb) is supplied to the comparators 12 and 44 for being compared respectively with the slope-compensated error signal V_(err2) and the threshold signal V_(peak) _(—) _(TH).

With the function of the comparator 12, the first comparison result signal CR1 immediately changes from the LOW level to the HIGH level once the current feedback signal V_(ifb) linearly increases in excess of the slope-compensated error signal V_(err2). On the other hand, with the function of the comparator 44, the second comparison result signal CR2 immediately changes from the LOW level to the HIGH level once the current feedback signal V_(ifb) linearly increases in excess of the threshold signal V_(peak) _(—) _(TH). The first and second comparison result signals CR1 and CR2 are coupled through the NAND gate 43 for controlling the reset terminal R of the SR latch 41. Therefore, the reset terminal R is triggered (i.e. the output signal of the NAND gate 43 changes from the HIGH level to the LOW level) under a condition that both of the first and second comparison result signals CR1 and CR2 must have the HIGH level. In other words, when the current feedback signal V_(ifb) is below the threshold signal V_(peak) _(—) _(TH), the output signal of the NAND gate 43 is kept HIGH and incapable of triggering the reset terminal R due to the LOW level of the second comparison result signal CR2, even if the current feedback signal V_(ifb) has already linearly increased in excess of the slope-compensated error signal V_(err2) to trigger the first comparison result signal CR1 from the LOW level to the HIGH level. As a result, the high-side switch HS stays conductive for allowing the current feedback signal V_(ifb) to linearly increase in excess of the threshold signal V_(peak) _(—) _(TH).

When the reset terminal R is triggered the inverting output terminal {overscore (Q)} of the SR latch 41 supplies the high-side drive signal HD and low-side drive signal LD with the HIGH level. The high-side drive signal HD with the HIGH level turns off the high-side switch HS while the low-side drive signal LD with the HIGH level turns on the low-side switch LS. As a result, the energy stored in the inductor L delivers to the load R_(L) for linearly decreasing the inductor current I_(L). Since the peak current flowing th rough the inductor L must become higher than or equal to the threshold peak current I_(peak) _(—) _(TH) for triggering the reset terminal R, the inductor current I_(L) are effectively prevented from decreasing to reverse polarity before the pulse signal PU triggers the set terminal S again to start the next switch cycle. Therefore, the synchronous switching buck regulator according to the present invention operates without a discontinuous mode and avoids the problems caused by the inductor current reversal and/or the high-frequency fluctuation of the output voltage.

In the embodiment shown in FIG. 4, since the comparators 12 and 44 are designed as the voltage-type comparators for receiving the voltage-type input signals and supplying the voltage-type output signals, the current feedback circuit 16 and the threshold peak current setting circuit 45 are designed to output the voltage-type signals V_(ifb) and V_(peak) _(—) _(TH) for indirectly corresponding to the physical quantity I_(L) and I_(peak) _(—) _(TH), instead of directly outputting current-type signals. It should be noted that the present invention is applicable to a case where the current feedback circuit 16 and the threshold peak current setting circuit 45 are designed to directly output the current-type signals. For example, current-to-voltage converters are additionally provided to the output terminals of the current feedback circuit 16 and the threshold peak current setting circuit 45 for converting the current-type signals to the voltage-type signals. Alternatively, voltage-to-current converter is employed to convert the voltage-type error signal V_(err2) to the current-type signal while the comparators 12 and 44 are designed as the current-type comparator circuits. In this case, the current feedback circuit 16 and the threshold peak current setting circuit 45 may be designed to directly output the current-type signals.

As clearly seen from the equation (1), the threshold signal V_(peak) _(—) _(TH) is a parabolic function of the duty ratio. Except for the establishment of the steady state, the duty ratio varies in accordance with the real-time operation of the switching voltage regulator. Even in the steady state the duty ratio may need to increase for maintaining the output voltage V_(out) at the desired value due to a drop of the input voltage source V_(in). Therefore, the threshold peak current setting circuit 45 according to the present invention must adjust the threshold signal V_(peak) _(—) _(TH) in accordance with the real-time operation of the switching voltage regulator, rather than generates a fixed signal to serve as the threshold signal V_(peak) _(—) _(TH).

FIG. 5 is a detailed circuit diagram showing a first example of a threshold peak current setting circuit 45-1 according to the present invention. As shown in FIG. 5, in response to the unregulated input voltage Vin and the ramp signal RA having the period T_(S), the threshold peak current setting circuit 45-1 generates the threshold signal V_(peak) _(—) _(TH) representative of the threshold peak current I_(peak) _(—) _(TH) calculated in accordance with the equation (1) described above. More specifically, the input voltage V_(in) is linearly amplified through operating amplifiers OP1 and OP2 in sequence to form the threshold signal V_(peak) _(—) _(TH), which may be expressed as the following equation (2): $\begin{matrix} {V_{peak\_ TH} = {\left( \frac{R_{v\quad 1}}{R_{c\quad 1}} \right) \cdot \left( \frac{R_{v\quad 2}}{R_{c\quad 2}} \right) \cdot V_{i\quad n}}} & (2) \end{matrix}$

wherein R_(c1) is a constant resistance connected between the input voltage V_(in) and an inverting input terminal of the operating amplifier OP1, R_(v1) is a linearly variable resistance connected between the inverting input terminal and an output terminal of the operating amplifier OP1, R_(c2) is a constant resistance connected between the input voltage V_(in) and an inverting input terminal of the operating amplifier OP2, and R_(v2) is a linearly variable resistance connected between the inverting input terminal and an output terminal of the operating amplifier OP2.

The linearly variable resistance R_(v1) is designed as a time-argument function that may be expressed as the following equation (3): R _(v1)(t)=R _(v1,t=0) ·D(t)  (3)

wherein R_(v1,t=0) is an initial value of the linearly variable resistance R_(v1) in the beginning of the switch cycle and D(t) is a time-argument function whose value linearly increases from zero to one over the period T_(S). The linearly variable resistance R_(v2) is designed as another time-argument function that may be expressed as the following equation (4): R _(v2)(t)=R _(v2,t=0)·(1−D)(t)  (4)

wherein R_(v2,t=0) is an initial value of the linearly variable resistance R_(v2) in the beginning of the switch cycle and (1−D)(t) is a time-argument function whose value linearly decreases from one to zero over the period T_(S). The oscillating circuit 11 shown in FIG. 4 generates the ramp signal RA having amplitude that linearly increases over the period T_(S). Consequently, the linearly variable resistance R_(v1) may be modulated in response to the ramp signal RA generated from the oscillating circuit 11. On the other hand, the ramp signal RA becomes an inverted waveform through an inverter INV to cause its amplitude to linearly decrease over the period Ts. Consequently, the linearly variable resistance R_(v2) may be modulated in response to the inverted ramp signal.

By substituting the equations (3) and (4) for the equation (2), the threshold signal V_(peak) _(—) _(TH) may be expressed as the following equation (5): $\begin{matrix} {{V_{peak\_ TH}(t)} = {{\left( \frac{R_{{v\quad 1},{t = 0}}}{R_{c\quad 1}} \right) \cdot \left( \frac{R_{{v\quad 2},{t = 0}}}{R_{c\quad 2}} \right) \cdot {D(t)} \cdot \left( {1 - D} \right)}{(t) \cdot V_{i\quad n}}}} & (5) \end{matrix}$

Comparing the equations (1) and (5), it is found that the proportional coefficient made up of the resistances must be designed to satisfy the following condition (6): $\begin{matrix} {{\left( \frac{R_{{v\quad 1},{t = 0}}}{R_{c\quad 1}} \right) \cdot \left( \frac{R_{{v\quad 2},{t = 0}}}{R_{c\quad 2}} \right)} = \frac{T_{s}}{L}} & (6) \end{matrix}$

During each switch cycle, the threshold signal V_(peak) _(—) _(TH) corresponding to this very moment when the current feedback signal V_(ifb) reaches the slope-compensated error signal V_(err2) is precisely generated from the threshold peak current setting circuit 45-1. As a result, the comparator 44 effectively determines whether the current feedback signal V_(ifb) reaches the threshold signal V_(peak) _(—) _(TH) or not, thereby avoiding operating the switching voltage regulator with the discontinuous mode.

FIG. 6 is a detailed circuit diagram showing a second example of a threshold peak current setting circuit 45-2 according to the present invention. As shown in FIG. 6, the input voltage V_(in) determines a current I_(a) through a linear current regulator constructed of an operating amplifier OP_(a), an NMOS transistor N1, and a resistance R_(a), which may be expressed as the following equation (7): $\begin{matrix} {I_{a} = \left( \frac{V_{i\quad n}}{R_{a}} \right)} & (7) \end{matrix}$

In other words, the current I_(a) is proportional to the input voltage V_(in). PMOS transistors P1 to P4 form a current mirror with the multiple output stages P2 to P4 for supplying the current I_(a). The transistor P2 supplies the current I_(a) to a differential pair constructed of PMOS transistors P5 and P6. The transistor P3 supplies the current I_(a) to another differential pair constructed of PMOS transistors P7 and P8. The transistor P5 has a gate electrode controlled by a low-boundary reference voltage V_(bl) while the transistor P8 has a gate electrode controlled by a high-boundary reference voltage V_(bh). The transistors P6 and P7 have their gate electrodes connected together for receiving the ramp signal RA generated from the oscillating circuit 11.

The current I_(a) supplied from the transistor P3 is distributed to the transistor P5 in accordance with a difference between the low-boundary reference voltage V_(bl) and the ramp signal RA. In other words, the differential pair constructed of the transistors P5 and P6 is controlled by the low-boundary reference voltage V_(bl) and the ramp signal RA for allowing a periodically-varying component of the current I_(a) to flow through the transistor P5 in accordance with the variation of the ramp signal RA. The current I_(a) supplied from the transistor P4 is distributed to the transistor P8 in accordance with a difference between the ramp signal RA and the high-boundary reference voltage V_(bh). In other words, the differential pair constructed of the transistors P7 and P8 is controlled by the ramp signal RA and the high-boundary reference voltage V_(bh) for allowing a periodically-varying component of the current I_(a) to flow through the transistor P8 in accordance with the variation of the ramp signal RA. Subsequently, the periodically-varying components flowing through the transistors P5 or P8 are combined together and then converted to a current I_(b) by a current mirror constructed of NMOS transistors N3 and N4.

The threshold signal V_(peak) _(—) _(TH) is implemented by a potential difference across a resistance Rb through which the currents I_(a), I_(b), and I_(c) flows, which may be expressed as the following equation (8): V _(peak) _(—) _(TH)=(I _(a) −I _(b) +I _(c))·R _(b)  (8)

The current I_(a) is generated to approximately simulate the variation of the input voltage V_(in) in the equation (1). Since the current I_(b) changes in response to the current I_(a) and the ramp signal RA, the current I_(b) is suitable for approximately simulating the variation of the threshold signal V_(peak) _(—) _(TH) caused by the duty ratio D and the input voltage V_(in). The current I_(c) is a constant offset current for shifting a DC level of the threshold signal V_(peak) _(—) _(TH). In one embodiment of the present invention, the low-boundary reference voltage V_(bl) is set at 0.5 volts, the high-boundary reference voltage V_(bh) is set at 0.75 volts, and the ramp signal RA is set to linearly change from 0 volt to 0.8 volts. Under such parameters, the combination of the currents I_(a), I_(b), and I_(c) expressed in the equation (8) is able to approximately simulate the threshold signal V_(peak) _(—) _(TH) under the situation that the duty ratio D is between 0.66 and 1.

It is should be noted that the circuitry and method according to the present invention is not limited to the preferred embodiment described above and can be widely applied to various topologies of switching voltage regulators, such as synchronous or non-synchronous, boost or buck, voltage feedback control or current feedback control, pulse width modulation (PWM) or pulse frequency modulation (PFM), and the like. FIG. 7 is a circuit block diagram showing a synchronous switching boost regulator according to the present invention. The boost regulator of FIG. 7 is different from the buck regulator of FIG. 4 in the connection relationship between the high-side switch HS and the inductor L, a switching logic circuit 70, and a threshold peak current setting circuit 75. As shown in FIG. 7, the inductor L is connected between the input voltage V_(in) and the common node CN while the high-side switch HS is connected between the common node CN and the output terminal. The switching logic circuit 70 supplies the high-side drive signal HD and the low-side drive signal LD through the normal output terminal Q of the SR latch. The threshold peak current setting circuit 75 generates the threshold signal V_(peak) _(—) _(TH) representative of the threshold peak current I_(peak) _(—) _(TH) flowing through the inductor L, which may be expressed as the following equation (9): $\begin{matrix} {I_{peak\_ TH} = {\left( \frac{T_{s}}{L} \right) \cdot D \cdot \left( {1 - D} \right) \cdot V_{out}}} & (9) \end{matrix}$

Comparing the equations (9) and (1), it is found that the output voltage term V_(out) of the equation (9) corresponds to the input voltage term V_(in) of the equation (1). Therefore, the threshold peak current setting circuit 75 applied to the boost regulator determines the threshold signal V_(peak) _(—) _(TH) in response to the output voltage V_(out) and the ramp signal RA generated from the oscillating circuit 11. More specifically, it is easy to obtain the threshold peak current setting circuit 75 applicable to the boost regulator of FIG. 7 if the output voltage V_(out) is substituted for the input voltage V_(in) of the threshold peak current setting circuit 45-1 shown in FIG. 5 or the threshold peak current setting circuit 45-2.

While the invention has been described by way of examples and in terms of preferred embodiments, it is to be understood that the invention is not limited to the disclosed embodiments. To the contrary, it is intended to cover various modifications. Therefore, the scope of the appended claims should be accorded the broadest interpretation so as to encompass all such modifications. 

1. A switching voltage regulator comprising: a switching circuit operating with a first operating state and a second operating state such that the switching circuit allows a switching current to linearly increase during the first operating state and allows the switching current to linearly decrease during the second operating state; a control circuit coupled to the switching circuit for controlling the switching circuit to operate with the first operating state or the second operating state; and a setting circuit for generating a threshold signal such that in response to the threshold signal the control circuit ensures that the switching current linearly increases to become higher than or equal to a current value set by the threshold signal during the first operating state, thereby preventing the switching current from linearly decreasing to reverse polarity during the second operating state.
 2. The switching voltage regulator according to claim 1, further comprising: an inductor connected to the switching circuit such that the switching current flows through the inductor.
 3. The switching voltage regulator according to claim 2, further comprising: an input voltage source for supplying energy to the inductor to linearly increase the switching current during the first operating state.
 4. The switching voltage regulator according to claim 1, wherein: the control circuit comprises: a current feedback circuit for generating a current feedback signal representative of the switching current, and a comparing circuit for comparing the current feedback signal and the threshold signal such that the control circuit prevents the switching circuit from operating with the second operating state when the current feedback signal is lower than the threshold signal.
 5. The switching voltage regulator according to claim 1, wherein: the setting circuit adjusts the threshold signal in response to at least a duty ratio of the switching circuit.
 6. The switching voltage regulator according to claim 1, wherein: the setting circuit adjusts the threshold signal in response to at least a period of time spent by the switching circuit operating with the first operating state.
 7. The switching voltage regulator according to claim 1, wherein: the switching voltage regulator converts an input voltage to an output voltage smaller than the input voltage, and the setting circuit adjusts the threshold signal in response to at least the input voltage.
 8. The switching voltage regulator according to claim 1, wherein: the switching voltage regulator converts an input voltage to an output voltage larger than the input voltage, and the setting circuit adjusts the threshold signal in response to at least the output voltage.
 9. The switching voltage regulator according to claim 1, wherein: the switching voltage regulator converts an input voltage to an output voltage, and the setting circuit comprises: a linear current regulator for generating a current signal representative of the input voltage; a first differential pair controlled by a periodic signal and a low-boundary reference voltage signal for selecting a first component from the current signal; a second differential pair controlled by the periodic signal and a high-boundary reference voltage signal for selecting a second component from the current signal; and a constant current source for supplying an offset current signal, wherein: the threshold signal is approximately simulated by a combination of the current signal, the first component, the second component, and the offset current signal.
 10. The switching voltage regulator according to claim 9, further comprising: an oscillating circuit for generating a pulse signal and a ramp signal such that the control circuit in response to the pulse signal controls the switching circuit to operate with the first operating state, and the periodic signal is implemented by the ramp signal.
 11. The switching voltage regulator according to claim 1, further comprising: an oscillating circuit for generating a pulse signal and a ramp signal such that the control circuit in response to the pulse signal controls the switching circuit to operate with the first operating state, and the setting circuit in response to the ramp signal adjusts the threshold signal.
 12. A method of controlling a switching voltage regulator, comprising steps of: controlling a switching circuit to operate with a first operating state for allowing a switching current to linearly increase, and controlling the switching circuit to operate with a second operating state for allowing the switching current to linearly decrease, wherein: in the step of controlling the switching circuit to operate with the first operating state, the switching current is ensured to become higher than or equal to a threshold current, thereby preventing the switching current from linearly decreasing to reverse polarity in the step of controlling the switching circuit to operate with the second operating state.
 13. The method according to claim 12, further comprising steps of: comparing the switching current and the threshold current, and maintaining the switching circuit to operate with the first operating state when the switching current is lower than the threshold current.
 14. The method according to claim 12, wherein: the threshold current is adjusted in accordance with at least a period of time spent for the step of controlling the switching circuit to operate with the first operating state.
 15. The method according to claim 12, wherein: the switching voltage regulator converts an input voltage to an output voltage smaller than the input voltage, and the threshold current is adjusted in accordance with at least the input voltage.
 16. The method according to claim 12, wherein: the switching voltage regulator converts an input voltage to an output voltage larger than the input voltage, and the threshold current is adjusted in accordance with at least the output voltage.
 17. A threshold current setting circuit comprising: a first circuit for generating a first current signal in response to a voltage signal such that the first current signal is proportional to the voltage signal; a second circuit for generating a second current signal in response to the first current signal and a periodic signal such that the second current signal is representative of a periodically-varying current; and a third circuit for generating a third current signal representative of a predetermined current value, wherein: a combination of the first to the third current signals is used to approximately simulate a threshold current signal.
 18. The circuit according to claim 17, wherein: the first circuit is implemented by a linear current regulator.
 19. The circuit according to claim 17, wherein: the second circuit comprises: a first differential pair controlled by the periodic signal and a low-boundary reference voltage signal for selecting a first periodically-varying component from the first current signal in accordance with a variation of the periodic signal, and a second differential pair controlled by the periodic signal and a high-boundary reference voltage signal for selecting a second periodically-varying component from the first current signal in accordance with the variation of the periodic signal, such that: the second current signal is formed by a sum of the first and the second periodically-varying components.
 20. The circuit according to claim 17, wherein: the third current signal is used to adjust a DC level of the threshold current signal. 